tional plane Figure 7(b) indicates diminishing pattern change directionality with increased board length. As board length increases, overall pattern differences addressed in Figure 8(a) decrease in the azimuthal plane.Although co-polarized pattern components are the focus of this work, cross-polarized components add insight to the pattern-difference dependence upon antenna geometry. For the two boards having 39-and 47-mm lengths and 79-mm width, the viewpoint inclusive of both polarizations reveals another aspect of handsetfeed interaction. The increased cross-polarized magnitude [see Figure 7(a)}, through the associated change in polarization state, indicates physical re-distribution of current on the ground plane. DISCUSSIONAntenna impedance sensitivity to board geometry was demonstrated through results presented for two handset designs. For electrically smaller handsets significant response changes between measurements including either the sleeve or regular transmission line devices in the RF feed were observed. The offset between the designed and measured operating band comprises a related effect. If the antenna-handset radiator is idealized to be a half-wave dipole, calculations showed plausibility of a resonance shift from the intended 1.9-GHz operating frequency to 1.75 GHz. Because the system consists of conductors of dissimilar geometry, length compensation techniques were not employed during design. Uncertainties introduced by application of the dipole analogy to an asymmetric antenna prompted dimension choices referenced to free-space wavelength. The match observed at 1.6 GHz suggests that designers should account for board size while tailoring the antenna impedance.Pattern difference mean and standard deviation were utilized to quantify pattern effects introduced by feed-line radiation. The statistical parameters help to contrast measurements of an isolated handset with those that include detrimental feed-line contributions. The mean and standard deviation together capture the isolation device's overall magnitude and direction-dependent changes to the radiation pattern shape. As such, complete assessment of pattern effects requires both and parameters. Electrically longer handsets show decreased sensitivity to cable radiation suppression techniques, while dimensions less than 0 /2 require careful design consideration. Because the 79-mm-width handset's measured and values dominate those corresponding to the 50-mm-width design, geometry and wavelength determine the extent of interaction between antenna and transmission line. ACKNOWLEDGMENTDielectric substrates were provided by Rogers Corporation. AN EQUIVALENT WAVEGUIDE APPROACH TO DESIGNING OF REFLECT ARRAYS WITH THE USE OF VARIABLE-SIZE MICROSTRIP PATCHES

The article provides an extensive review on the topic of microwave and millimeter‐wave power combiners and dividers. Because many power combining and dividing structures can be used interchangeably, the review focuses on power combiners. It commences with an introduction of power combining principles and grouping of power combiners into suitable categories. The main categories considered here include circuit‐level (both resonant and nonresonant type) and space‐level power combiners. Next, operation and fundamental characteristics of circuit and space‐level combiners, which have been presented in the microwave literature, are thoroughly explained. In the subsequent sections, analysis methods for power combiner/divider structures are described. They concern both circuit and electromagnetic field approaches. The use of state‐of‐the art computer software for the analysis of power combining/dividing structures is also covered. The last section of the article is concerned with applications of power combiner/dividers in modern communication and radar systems. Examples of most impressive power combiners of solid‐state devices, in terms of an achieved output power level or highest frequency of operation, are presented. The review concludes with remarks on recent progress in the field of space‐level combiners.

respectively, for the sample frequencies 7 GHz and 8 GHz where the main lobes are along -y direction in Figure 1 for both patterns. According to the results in these figures, the patterns are said to be symmetric and almost Gaussian. Besides, the patterns are quite insensitive to frequency changes in the given operation frequency band especially for H-plane that the patterns do not show crucial changes even for 1 GHz change in the frequency. The gains of the sole dielectric antenna are 13.1 dBi and 14 dBi for 7 GHz and 8 GHz, respectively where the gain of the antenna varies from 12.7 to 14.3 dBi within the frequency band 6.7-8.25 GHz.The cross-polarization characteristics, which is an another important fact for an antenna, are also examined for the given antenna. For this purpose, the copolarization and cross-polarization patterns of the antenna corresponding to E-plane and Hplane are given in Figure 11 and 12, respectively, for the sample frequency 7.5 GHz. According to these patterns belonging to 7.5 GHz, whereas the cross-polarization level is about 27 dB below copolarization level within the elevation angle range 90 6 30 (60-120) for E-plane, this level is about 30 dB below for the azimuth angle range 270 6 90 and about 40 dB below for the azimuth angle range 270 6 30 for H-plane. Throughout the frequency band 6.7-8.25 GHz, the relative cross-polarization level is at most À25 dB for elevation range 90 6 30 and À35 dB for azimuth range 270 6 30 . Therefore, a low cross-polarization system can be achieved by designing a proper parabolic reflector antenna, which takes the reflected fields of the proposed antenna within 30 beamwidth in both principal planes into account. CONCLUSION

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