A balanced quad‐band diplexer is proposed with wide common‐mode (CM) suppression and high differential‐mode (DM) isolation in this study. The six‐port balanced diplexer was formed by combining two dual‐band bandpass filters operating at 1.92/5.25 and 2.45/5.8 GHz through two stepped‐impedance resonator (SIR) inserted T‐junctions. By carefully designing the section length of the T‐junction and the SIR length from its open‐end to the joint‐point with the T‐junction, high isolation between diplexer channels can be attained. A prototype of the proposed balanced quad‐band diplexer was implemented and measured with a DM isolation of higher than 51.8 dB and a CM suppression of larger than 47.9 dB in the four DM passbands. Good agreement was observed between the simulated and the measured results.

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normal incidence, the measured resonant frequency in the Xband appears at 8.2 and 8.9 GHz. The difference between the measured and simulated results for the K-band can be considered as cable losses, and the gap between the DFSS with the infinite arrays for computer simulation, and the DFSS with finite arrays for measurement. However, in Figure 12, it can be clearly seen that the measured and simulated results are in good agreement, and can be accepted. CONCLUSIONIn this article, to obtain angle and polarization insensitivities and broad bandwidth, a DFSS is proposed, composed of ring patch elements of four different sizes. To enlarge the bandwidth, and to eliminate the interruptions of the transmission peaks, the thickness of foam layer between the FSS layers is adjusted. The proposed DFSS reflects the X-and K-band signals, while transmitting through the C-and Ku-band signals. It is interesting to note that the resonances for X-and K-bands are stable, about variations of incident angle and thickness of the foam. To prove verification of the proposed DFSS, the proposed DFSS is fabricated and measured. The measured transmission coefficients for the proposed DFSS are compared with the simulated ones, and the measured and simulated results are in good agreement. ABSTRACT: A Q-band bidirectional transceiver has been designed and fabricated for millimeter-wave CMOS phase array systems. The proposed design replaces several switches previously required for a bidirectional approach with a compact double-pole double-throw switch. Phase shifting is performed in 90 step using a high-pass/low-pass structure, and the transceiver demonstrates a highly linear phase response. Over the frequency of 35 to 40 GHz, the measured root-mean-square phase error is <5.5 and <12.1 for transmitter and receiver paths, respectively. Key words: CMOS; millimetermillimete-wave; phase array; phase shifter; switch INTRODUCTIONA phase array transceiver requires separate receive (Rx) and transmit (Tx) channels. The size of the phase array system can be reduced significantly by sharing a passive phase shifter for Tx and Rx operation [1]. One disadvantage of the conventional bidirectional approach is the additional switches required at the Tx and Rx outputs. This article discusses the phase and gain response achieved using a bidirectional two-bit transceiver for Q-band phase array system. Several switches required for bidirectional function are replaced with a new double-pole doublethrow (DPDT) switch. Because the size of DPDT switch is very small, the proposed approach allows a simple and compact realization of the phase array transceiver. Furthermore, good isolation of the new approach achieves a good phase response, where root-mean-square (rms) phase error is <5.5 and < 12.1 for transmitter and receiver paths, respectively, over the broad bandwidth of 35 to 40 GHz. SYSTEM AND CIRCUIT DESIGNA schematic for the bidirectional transceiver is shown in Figure 1. The phase shifter uses a high-pass/low-pass configuration which is suitable for dense phase...

patch will be kept unchanged as long as the cutting plane can be viewed magnetic wall. Under such situation, HCSRR, sharing the same magnetic wall with the half patch antenna, can be suitably sandwiched between the radiating patch and the ground plane, thus, a further miniaturization can be realized. Given that the electric field of the dominant mode within a patch cavity is kept unchanged, it is possible to excite properly designed HCSRR sandwiched between the ground plane and radiating patch and, thus, lower the resonant frequency of the half patch.Comparison of the return losses among four different types antennas are made in Figure 5. To make the explanation concise, circular patch antenna, circular patch antenna loaded with CSRR, half circular patch antenna, and half circular patch antenna loaded with HCSRR are renamed as Type-A, Type-B, Type-C, and Type-D antenna, respectively. As depicted in Figure 5, in terms of the fundamental mode, it can be observed that Type A and Type C antennas almost resonate at the same frequency (near 4 GHz), while Type A and Type C antennas almost resonate at another same frequency (near 2.2 GHz). It can be concluded that the half beveled technology shows little impact on the impedance matching of the antenna. Moreover, when the HCSRR (CSRR) is loaded, the resonant frequency decreases to 2.2 GHz, which is much lower than the original 4 GHz. Through the above analysis, conclusion can be drawn that HCSRR can reduce the antenna size to half while the impedance characteristic is kept almost unchanged. By the way, weaker resonances occurring at around 3.3 and 3.75 GHz mainly result from the higher order resonance of the HCSRR.To validate the above design, a prototype of the proposed antenna is fabricated on the two 1 mm-thickness Rogers RT/ duroid 5880 substrates (e r 5 2.2) with the parameters optimized by the simulation software HFSS. The photograph of the proposed antenna is depicted in Figure 6 and the parameters of the antenna structure are listed in Table 1. As shown in Figure 7, the fabricated antenna operates at 2.26 GHz, which is about 0.05 GHz higher than the simulated result. Figure 8 depicts the radiation patterns for the proposed antenna operating at 2.26 GHz. Good accordance between the simulated and measured results is obtained. The measured copolarization is at least 10 dB higher than the cross polarization in terms of the broadside radiation. As stated above, these results can effectively validate the characteristic of the proposed HCSRR structure. Moreover, a further size reduction for the HCSRR can be realized through introducing more slots, which owns the similar working principle as the one-slot HCSRR structure. CONCLUSIONA novel miniaturized HCSRR structure is investigated and its employment for designing miniaturized patch antenna is researched in this letter. The HCSRR shows almost the same resonant characteristic as the original CSRR, which can be explained in terms of the equivalent circuit model. Then the circular patch antenna can be beveled into a half an...